Transcript 80GHz Modulator Designs Ian Harrison School of Electrical and Electronic Engineering University of Nottingham UK Work done at Department of ECE University of California, Santa Barbara USA Special thanks PK,

80GHz Modulator Designs
Ian Harrison
School of Electrical and Electronic Engineering
University of Nottingham
UK
Work done at
Department of ECE
University of California, Santa Barbara
USA
Special thanks
PK, Zak, Mattias for fabrication of circuits and devices
Miguel for advice
Paidi and Navin for cricket discussions
Mark Rodwell for useful discussion and use of infrastructure
[email protected] 805-893-8044, 805-893-3262 fax
Introduction
• Concentrate on more recent work
• Thermal Modelling
• Modulator work
– design issues
– Simulation results
Design Specifications
• Two types of optical modulator
– LiNb03 Mach Zehnder -Interference
• Split beam into 2, induce 0 or 180 phase shift
• Large driving voltage eg 10GBits 5Vpp
– Electroabsorption
• Quantum confined stark effect
• Smaller driving voltage eg 10GBits 3Vpp
EA modulator
2V , 50 Ohm input
Output should be matched
E=0
Attn
• Design specifications
E≠0
E=0
E≠0
λ
How do we get speed improvement
• Switching speed limited by output capacitance
C.V

I
Design Specifications set ΔV
and RL  sets I
Formula simplistic
insight
Reduce C by decreasing AC
 Increase in J since I fixed
 J limited by Kirk Effect
 Increase in J increase dissipated power density
Kirk effect and switching time
Above Jkirk massive increase in base charge
 Base push out (Field Screening)
J Kirk
(Vcb   )4vsat

2
TC
AC V TC

AE Vce 4vsat
Wide emitter, narrow base mesa
Rb limits the emitter width
MAX
VCE
Predicts
straight line
J
•
(mA/m)
3.5
3
Vsat=3.5 105cms-1
2.5
2
1.5
0.6
0.8
1
1.2 1.4
V (V)
CE
1.6
1.8
Why is thermal management important?
•
As J increases so does the power
density. This will lead to an
increase in the temperature.
TC
JKirk
Le
Å
mAμm-2
μm
3000
1.0
81
2000
2.3
34
1500
4.1
19
1000
9.8
8.6


V=2V
80mA
For VCE=1V  PD=10.6mWμm-3
For VCE=1V  PD=98mWμm-3!!
Thermal Modeling of HBT (1)
• 3D Finite Element using Ansys 5.7
• K (Thermal conductivity) depends temperature
• K depends on doping
 300 
kT  k300 

 T 
n
• For GaAs heavily doped GaAs 65% less than undoped GaAs
• Unknown for InP or InGaAs use GaAs dependency
Material
K300
n
K300(exp)
Refs
InP
0.68
1.42
0.68-0.877
1
InGaAs
0.048
1.375
0.048-0.061
2
Au
3.17
-

Large uncertainty
in values
3
J.C.Brice in “Properties of Indium phosphide” eds S Adachi and J.Brice pubs INSPEC London p20-21
S Adachi in “Properties of Latticed –Matched and strained Indium Gallium Arsenide” ed P Bhattacharya pubs INSPEC London p34-39
“CRC Materials science and engineering handbook”, 2nd edition ,eds J.F Shackelford,A.Alexander, and J.S Park, pubs CRC press, Boca Raton, p270
Layout used for simulation validation
Need simplified model for simulation

Actual device
reduce simulation time and storage requirements
Ignore base pad collector interconnect
•2 orthogonal symmetry lines
•Simulate only ¼ device
Layer structure
Emitter
Base
Collector
Setback
Grade
Drift
Subcollector
Etchstop
Substrate
0.04μm n+ InGaAs
0.12 μm n- InP
0.03 μm p+ InGaAs
Polyimide for
passivation
Very low K ignore
In thermal analysis
After M. Dahlstrom
4
0.02 μm InGaAs
0.024 μm Grade
0.156 μm InP
1.5
7.2
3.5
4
2
1
14
0.7
0.050 μm n+ InGaAs
0.200 μm n+ InP
500 μm Fe: InP
2.5
0.25
0.1
102.5
Simulated ¼ Device
4
0.5
102.5
500
Validation of Model
40
Caused by
Low K
of InGaAs
Max T in
Collector
Temperature Rise (K)
35
center
Edge
30
25
20
15
10
5
SC
ES
C
B
E
E Metal
0
-0.2
0
0.2
0.4
0.6
0.8
1
1.2
Distance from substrate (m)
Ave Tj (Base-Emitter) =26.20°C
Measured Tj=26°C
Good agreement.
Advice
Limit InGaAs
Increase size of emitter arm
Effect of decreasing collector thickness
Assumptions
Devices thermally isolated

Device structure identical to
validation structure
Choose Le
For J=JKirk
We=0.5um
Perfect switching waveform
50% duty
cycle
Increase in ISC possible failure mechanism
( Major failure problem in GaAs HBT’s)
Temperature of one device approximately
double when circuit is not switching
6mA
360
12
350
10
340
8
330
6
320
4
310
2
0
300
0.05 0.1 0.15 0.2 0.25 0.3 0.35 0.4 0.45
Collector Thickness (m)
Length of Emitter (m)
Collector temperature always higher than Tj
(ΔTMax-ΔTj)>30°C )
Temperature (K)
Observations
Temperature increases rapidly for thin
collectors (ΔTmax =60°C for TC=1000Å)

V=0.3V
Analysis of 40,80,160 Gbit/s devices
•
To obtain speed inprovements require to scale other device
.
parameters
Speed
(Gbit/s)
40
80
160
Collector Thickness
(Å)
3000
2000
1000
Base Sheet resistance
()
750
700
700
Base contact resistance
(-m2)
150
20
10
Base Thickness
(Å)
400
300
250
Base Mesa width
( m)
3
1.6
0.4
Current Density
(mA/m2)
1
2.3
9.8
Emitter. Junction Width
( m)
1
0.8
0.2
Emitter Parasitic resistivity
(-m2)
50
20
5
Emitter Length
( m)
6
3.3
3.2
Predicted MS-DFF
(GHz)
62
125
237
Ft
(GHz)
170
260
500
Fmax
(GHz)
170
440
1000
Tj
(K)
7.5
14
28
TMax
(K)
10
20
49
TMax (No Etch Stop layer)
(K)
7.5
13
21
Device parameters after Rodwell et al


V=0.3V
6mA
Reduction of
parasitic CBC
Conservative
1.5x bit rate
When not switching
values will double
Thermal Analysis using ADS
R network easily solved
Using ADS
• For simulations need a model that can be solved by ADS so
that thermal and circuit simulations can be coupled.
•
•
•
•
Thermal generation

current source
Thermal resistance

resistors
Thermal capacity

capacitors (If static not needed)
Temperature variation of thermal conductivity not modelled because resistors do
not depend on current (This restriction could be lifted)
Coupled Circuit-Thermal modelling
•
How do the advance device
models do it?
– Device at one temperature
– Devices thermally isolated and
described by a single resistance
– Thermal circuit hidden from user
•
How do we want to do it
–
–
–
Access to thermal circuit
β only slightly temperature dependent
Large change in VBE(ON)
VBE    T
•Β is the band gap shrinkage factor
Not usually given but optical measurements on
band gap ( Optical values must be used with
caution )
0.0004 for both InP and InGaAs
Value used in model
Ambient T
Temperate rise
Power dissipated
in the device
Thermal
Resistance
My model
Can we measure Rth (Method of Lui et al )
0.010
Ramp IB for different VCE
Measure VBE and IC
IC.i
0.008
0.006
0.004
0.002
0.000
0.48
Depends on current density
0.54
VBE
5000
4000
0.52
Large uncertainty in values.
Fitting regression curves
helps to reduce error
RT
VBE
RT 
 VCE I C
0.50
3000
2000
1000
2
3
4
PAve
5
6
-3
x 10
An alternative method for finding RT
IC fixed , sweep VB
 Obtain RT (Pave)
 Changes in VC larger more accurate
 RT measured at lower Pave
Thermal instability possible
Need to be careful on the VB range
5000
2.5
Ic= 6mA,6mA
Ic=12mA,12mA
From gradient RT 4000
RT
2
3000
1.5
1 dVBE
RT 
 I C dVCE
1
0.5
0.74
RoT= 1945
0.76
0.78
0.8
0.82
2000
1000
0
Ic= 6mA,6mA
Ic=12mA,12mA
0.01
0.02
PIn
0.03
Comparison of the two methods
Emitter Mask 12 x 0.7 mesa width 1.7
“New method”
Classic Method
5000
Linear interpolation.
RT
3000
3000
2000
2000
1000
0
0.005
0.01
PAve
0.015
0.02
Empirical Curve fit
RT
4000
3000
2000
0.005
0.01
PAve
1000
0
0.01
0.02
0.03
PIn
5000
1000
0
RoT= 1945
4000
RT
4000
5000
0.015
0.02
Classic method badly affected by the 4145
resolution.
Better measurements at very high power.
Often leads to device failures
Problems with every fourth measurement of
4145 in “new” method
Need to compare the two methods using the
4155
Which model to estimate Rth
•
Finite elements clearly shows diffusion of heat
along the collector under the base contacts.
•
Rth should depend on base mesa size
Model 1
 Models flow of heat under base
 Thermal circuit complex
Model 2
 Thermal circuit simple
 Over estimates RT
Both Models
 Both will underestimate RT at high powers
Experimental results
Model 2
Mesa Width
Length
Model 1
RT(C/W) 1.7
2.1
2.7
4
5500
5100
6200
12
1800
 Use Model 2
1800
Thermal resistance calculations
•
Thermal resistance of layers can be
estimated from the thermal conductivity if
no heat spreading is assumed.
•
The emitter interconnect acts as a thermal
link
•
The thermal resistance of the substrate is
estimate by solving the 3D heat flow
problem using separable variables
technique. This is the same method Lui et
al used to calculate RT of single and multifinger HBT power transistors.†
Length
Mesa Width
RT(C/W)
1.7
2.1
2.7
Theory
4
5500
5100
6200
5700
12
1800
1800
2071
After M. Dahlstrom
Spreadsheet: ThermalCalc.xls
Stability of single BJT’s (Intro)
0.010
•
•
•
Well known problem solved by
ballasting with emitter or base
resistance.
Known to be a problem in power
amplifiers.
May argue, incorrectly, that in
digital circuits this is not a
problem because we are driving
the circuits with a constant current
source.
Need to know how large we can
make the emitters before “hot
spots” form and current “hogging”
becomes an issue.
0.008
IC.i
•
0.006
X
0.004
0.002
0.000
0.48
0.50
0.52
0.54
VBE
If the transistor base is being driven with
a constant voltage.
The collector current will increase until it
gets to point X.
Any further increase in base voltage will
cause an infinite increase in the collector
current resulting in physical damage to
the device.
Single Emitter Stability
VC
Max
qIC RE   kT

I C RT (q  k )
Caused by the increase in RT
when device size is reduced.
Uncertainty in Re
Maximum VC for different IC.
Emitter Length (Mask) =10m
2.4
0
3.4
3.2
Max
2.8
2.6
3.6
VC (V)
VMax
(V)
C
Stability curves against emitter length for const J
3.2
Caused by the reduction
of Re with length
3
-2
JC= 2 mAm
JC= 3 mAm-2
5
10
15
Emitter Length (Mask) (m)
Optimum operating point
Theory
Experiment
J =1  5mAμm-2
3
2.8
20
2.6
0
0.01
0.02
0.03
Collector Current (A)
ρE=60Ω from DC measurements
Hot spot formation (not finished)
Device broken into sections
Thermal model of substrate
Base electrical
resistance
Need to do
1. Simulate DC measurements
2. Compare with measurements
Thermal resistance
of the emitter
connection
Modulator design (Matching)
Passive
 simple  high bias current
Passive
All active circuits
Bias current lower  need to prevent saturation
Resistive feedback
No flexibility Zo=1/gm
Feedback
Zo=1/(gmβ) but additional EF more ringing
Resistive Feedback
RC
Feedback
Feedback
β<1
Effect of current source design on output
Current switch (only one half)
Vo
Vm
Capacitive coupling to
Control line reduces output resistance
Vi
RiseI
1.556E11
FallI
2.161E11
RiseO
3.366E11
FallO
8.172E11
RiseI
1.476E11
2.5
RiseO
3.577E11
FallO
6.014E11
2.5
Common
Reference
2.0
1.5
Different
Reference
2.0
Vo
1.5
1.0
1.0
Voltage
Voltage
FallI
2.012E11
0.5
0.5
0.0
0.0
-0.5
-0.5
Vm
-1.0
Vi
-1.5
-5
0
-1.0
-1.5
5
10
15
20
Time (ps)
25
30
35
40
-5
0
5
10
15
20
Time (ps)
Use resistor:- inefficient power use, but simple
25
30
35
40
Output stage options
Performance depends on
the quality of the ground
Bias generated
by diode
Miller effect increases output
cap
1.5
Ideal Vsrc
Output Voltage
1.0
With diode base
0.5
0.0
-0.5
-1.0
-1.5
-5
0
5
10
15
20
time, psec
25
30
35
40
Current designs
• 2 and 3 stage amplifiers
• Cascode and simple output
3 stage cascode output
80GBit/s
160GBit/s
1.5
1.5
1.0
Output Voltage
Output Voltage
1.0
0.5
0.0
-0.5
-1.0
0.5
0.0
-0.5
-1.0
-1.5
-5
0
5
10
15
20
time, psec
25
30
35
40
-1.5
-2
0
2
4
6
8
10
12
14
16
18
20
time, psec
Simulations show that 160GBit is just possible with 1500A
collector.
What to do in the future
•
•
•
•
Fabricate and test the current design
Design amplifiers with output voltage
Simulate with self heating
Investigate the more advanced BJT
models
Conclusion
• 160 Gbits Modulator has been designed
• Electro -thermal model has been
developed which can be simulated using
ADS
What would I change if I could rewind the
clock
Gone in the clean room.